Monitoring phase non-linearities in an optical communication system

ABSTRACT

Phase nonlinearities of an optical communications system are monitored by generating a test signal which includes a predetermined property that is uniquely associated with at least one phase nonlinearity of the optical communications system. The predetermined property of the test signal is then detected at a monitoring point of the optical communications system, and used to estimate the associated phase nonlinearity.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a divisional of patent application Ser. No.10/389,804, filed Mar. 18, 2003, entitled MONITORING PHASENON-LINEARITIES IN AN OPTICAL COMMUNICATIONS SYSTEM, the entirety ofwhich is incorporated herein by reference.

MICROFICHE APPENDIX

n/a

TECHNICAL FIELD

The present invention relates to optical communications systems, and inparticular to monitoring phase non-linearities in an opticalcommunications system.

BACKGROUND OF THE INVENTION

In modern optical communications networks, it is generally desirable totransmit optical signals at high power levels in order to maintainsufficient signal to noise ratios over extended transmission distances,and thereby obtain an acceptably low Bit Error Rate (BER) in a receivedoptical signal.

However, conventional optical waveguides (such as optical fibers)comprise an optical transmission medium which exhibits nonlinear effectsat high optical power levels, resulting in degradation of the opticalsignal. Nonlinear effects may similarly occur within optical terminalsof the system, in optical transmission media or in components such asoptical amplifiers. The optimum power level at which optical signals canbe transmitted is typically the maximum power level at which significantdegradation due to nonlinearity is avoided. Since the performance ofvarious optical components within the system varies with operatingconditions, age, and component replacement, a safety margin is used insetting the maximum power level. Consequently, optical communicationssystems typically operate at power levels which are less than theoptimum power level. A detailed discussion of nonlinear optical effectsis provided by Agrawal, Govind P., “Nonlinear Fiber Optics”, 2^(nd).Ed., Academic Press, Inc., San Diego, Calif., 1995 (ISBN 0-12-045142-5).

Of particular concern in considering nonlinear processes are the effectsof phase nonlinearities, which increase as data rates and optical powerlevels increase, and which ultimately limit both system performance andsignal reach.

Phase nonlinearities are the result of complex interactions between theoptical power present in the fiber; the refractive index of the fibermedium, including the non-linear index coefficient; the wavelengthdivision multiplexing (WDM) channel spacing; the polarization states ofthe signals within each of the channels; and the proximity of channelwavelengths to the zero-dispersion wavelength of the fiber. Phasenonlinearities include self-phase modulation (SPM), cross-phasemodulation (XPM), and modulation-instability (MI), all of which arediscussed in detail in Agrawal (supra), at chapters 4 and 7.

Self-phase modulation (SPM) is a by-product of the relationship betweenthe refractive index of the fiber medium and the optical power presentin the fiber. In particular, changing optical power causes a change inthe refractive index of the fiber medium. The refractive index change isproportional to the optical power level. Changing the refractive indexproduces a Doppler-like frequency shift (or chirp) that is proportionalto the time-rate of change of the refractive index (and, equivalently,the optical power level). Thus, changing optical power levels due tomodulation of an optical signal causes a frequency-shift (or chirp)within the signal itself. For example, consider an isolated signal pulse(e.g., an isolated binary “1”) launched into the optical fiber. SPMresults in the leading edge of the pulse being red-shifted (that is,frequency shifted toward the red end of the optical spectrum), and thetrailing edge of the pulse blue-shifted. Chromatic dispersion of thefiber will then cause these red- and blue-shifted portions of the pulseto propagate through the fiber at different speeds, which may result intime-domain distortion of the original pulse shape.

As may be appreciated, because the magnitude of the frequency shift isproportional to the time-rate of change of the optical power level, theamount of red- and blue-shift experienced by the pulse edges will be afunction of the rise and fall times at the leading and trailing edges,and the peak power level of the pulse. In additional to these factors,the total time-domain distortion experienced by the pulse will also beaffected by the nominal length of the pulse, and the length of the fiberbefore signal detection and regeneration. Clearly, the effects of SPMbecome increasingly severe as signal power, data rate (or spectralefficiency), and fiber span length are increased.

Cross-phase modulation (XPM) is similar to SPM, and produces the samefrequency-shifting effects, but occurs in Wavelength DivisionMultiplexed (WDM) systems. XPM is always accompanied by SPM, and occursbecause the effective refractive index “seen” by an optical wavepropagating in the fiber medium depends not only on the intensity ofthat wave but also on the intensity of other co-propagating waves. Thus,refractive index changes due to rising and falling optical power levelsin one channel induce corresponding frequency-domain distortions(chirps) within co-propagating signals (in adjacent channels). Chromaticdispersion of the fiber may then induce time-domain distortions of thosesignals, in the same manner as described above.

Modulation instability (MI) is an XPM-induced interaction betweenco-propagating optical waves (whether due to signal traffic, noise, orpump laser signals) within the optical fiber. This interaction producesnew, unwanted wavelengths (or side-bands) that can interfere with,and/or couple power from, desired optical signals.

Nonlinear effects in an optical fiber can be measured using knownoptical signal and spectrum analysis equipment. Respective channels of aWavelength Division Multiplexed (WDM) communications system can bemonitored, either by multiple signal analyzers arranged in parallel, orusing a single signal analyzer that is sequentially tuned to receiveeach optical channel signal in turn. Optical Spectrum Analyzers (OSAs)can be used to determine average and peak power levels, as a function ofwavelength, across a desired range of wavelengths. Known analyticaltechniques can be used to determine non-linear effects from the datameasured by these systems.

Due to their cost and complexity, conventional optical signal andspectrum analysis equipment is typically restricted to laboratory use.Furthermore, accurate measurement of nonlinear effects using suchequipment typically requires specialized test set-ups, which, again, canonly be provided in a laboratory setting.

In order to monitor nonlinearities in installed optical communicationssystems, simpler and less expensive monitoring equipment is required.Typical (in situ) optical performance monitoring systems known in theart are disclosed in co-assigned U.S. Pat. Nos. 5,513,024; 5,949,560;5,999,258; 6,128,111; 6,222,652; and 6,252,692. While these systemsenable some degree of performance monitoring, they tend to suffer anumber of disadvantages. In particular, per-channel monitoring systemsare typically dependent on a low frequency pilot tone (or dither) havingknown parameters. Any error between the design and actual parametervalues of the launched pilot tone will naturally degrade the accuracy ofany performance parameters calculated at the monitoring point.Additionally, this approach assumes that performance parameterscalculated on the basis of the low frequency pilot tone will be validfor the high-speed data traffic. Consequently, frequency-dependenteffects (most notably phase nonlinearities) cannot be detected with thisarrangement. Finally, the detectors and signal processors utilized inthese monitoring systems are low frequency analog devices. Thisprecludes their use for monitoring high-frequency phenomena such as SPM,XPM and MI.

Accordingly, a method and system that enables efficient monitoring ofphase nonlinearities in an installed optical communications systemremains highly desirable.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a method and system formonitoring phase nonlinearities in an optical communications system.

Accordingly, an aspect of the present invention provides a method ofmonitoring phase nonlinearities of an optical communications system.According to the present invention, a test signal is generated, whichincludes a predetermined property that is uniquely associated with atleast one phase nonlinearity of the optical communications system. Thepredetermined property of the test signal is then detected at amonitoring point of the optical communications system.

In some embodiments, the test signal is generated by simultaneouslytransmitting probe and data signals through respective channels of theoptical communications system. In this case, the probe signal has anarbitrary frequency, but has a rise and fall time that is selected toemulate that of live data traffic. This arrangement produces a testsignal, at the receiving end of the optical communications system, whichcomprises the data signal and a distortion component in the form offrequency chirps that are uniquely associated with cross-phasemodulation (XPM) between the probe and data signal channels. Thefrequency chirps in the test signal can be measured by a conventionalclock-recovery circuit, and isolated (from other frequency-domain noise)by correlation with rising and falling edges of the probe signal.Accuracy of this correlation function can be improved by prefilteringthe received probe signal prior to the correlation.

In other embodiments, the test signal is generated by imposing aselected dispersion to a data signal received through a selected channelof the optical communications system. For “flat portions” of the testsignal, the variation in signal noise (e.g. as represented by bit errorrate) with changes in the imposed dispersion can be readily evaluated,and is uniquely associated with non-linearity in the opticalcommunications system.

In further embodiments of the present invention, a respective high-speedanalog-to digital detector is used to sample each channel of the opticalcommunications system. The sample data is stored in association with thecorresponding recovered data generated by conventional data recoverycircuits. In this case, the test signal comprises an isolated signalpulse within the recovered data. A digital signal processor (DSP) canthen analyze the associated sample data to determine the correspondingsignal waveform. Changes in the signal wave form with changingtransmission power level of the involved channel provides a directmeasure of self-phase modulation (SPM). Correlation betweensimultaneously stored sample data of adjacent channels provides a directmeasure of cross-channel effects, such as XPM and cross-talk. Ifdesired, a probe signal as described above, can be used to simplify thecorrelation algorithm implemented in the DSP for evaluating XPM.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features and advantages of the present invention will becomeapparent from the following detailed description, taken in combinationwith the appended drawings, in which:

FIG. 1 is a block diagram schematically illustrating principal elementsof an optical communications network in which the present invention maybe deployed;

FIGS. 2 a-2 d schematically illustrate principal elements and operationsof a monitoring system in accordance with a first embodiment of thepresent invention;

FIG. 3 is a block diagram schematically illustrating principal elementsof a monitoring system in accordance with a second embodiment of thepresent invention;

FIGS. 4 a-4 c schematically illustrate principal operations of amonitoring system in accordance with a third embodiment of the presentinvention;

FIG. 5 is a block diagram schematically illustrating principal elementsof a data regenerator for implementing the monitoring system of FIGS. 4a-c; and

FIG. 6 is a block diagram schematically illustrating principal elementsand operations of a monitoring system in accordance with a fourthembodiment of the present invention.

It will be noted that throughout the appended drawings, like featuresare identified by like reference numerals.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention provides a method and system for monitoring phasenonlinearities, such as self-phase modulation (SPM), cross-phasemodulation (XPM), and modulation-instability (MI) in an opticaltransmission system. In general, a test signal having a predeterminedproperty that is uniquely associated with at least one type of phasenonlinearity is generated. The predetermined property of the test signalis detected at a selected monitoring point of the optical communicationssystem. For the purpose of the present description, the monitoring pointis located at a receiving node of an optical communications link.However, it will be appreciated that the monitoring point may be at anydesired location capable of receiving WDM signal traffic. The magnitudeof the phase nonlinearity is then evaluated from the detected value ofthe predetermined property. FIG. 1 is a block diagram schematicallyillustrating principal elements of an exemplary optical communicationssystem in which the present invention may be deployed.

As shown in FIG. 1, the optical communications system 2 comprises a pairof network nodes 4 a and 4 b interconnected by a link 6. The networknodes 4 may be any discrete electro-optical network devices (such as,for example, routers, add-drop multiplexers, etc.) known in the art.Similarly, the link 6 may be provided as a cascade of interconnectedoptical components (e.g., optical amplifiers interconnected by opticalfiber spans) in a manner well known in the art. For the purposes ofdescription of the present invention, the link 6 comprises an opticalmultiplexer 8, and an optical de-multiplexer 10, which areinterconnected by a single optical fiber span 12. However, it will beunderstood that the present invention is not limited to such embodiment.

In general, traffic in the optical communications system 2 may bebi-directional. However, in order to simplify description of theinvention, data traffic within the link 6 will be considered to beuni-directional, being launched through the link 6 by sending node 4 a,and propagating in a so-called “forward” direction 14 through the link 6to the receiving node 4 b.

FIGS. 2 a-2 d schematically illustrate principal elements and operationsof a monitoring system in accordance with a first embodiment of thepresent invention. The embodiment of FIG. 2 is particularly suited formonitoring cross phase modulation (XPM) between a pair of channels. Inthe illustrated example, XPM is monitored between a first channel, whichis nominally designated as channel A, and a second channel, nominallydesignated as Channel N. Thus a probe signal p(t) is transmitted throughchannel A using carrier wavelength 4, while a data signal D(t) issimultaneously transmitted through channel N using carrier wavelength4T. As the two signals co-propagate through the link 6, XPM betweenchannels A and N generates a test signal T(t) which is received, by thereceiving node 4 b, through channel N.

As is known in the art, XPM within the link 6 can be mathematicallymodeled (at least to a linear approximation) by means of a “transfer”function H(f); where the term (f) refers to the frequency variableacross the base-band modulation spectrum. With this formulation, theXPM-induced components in the test signal T(t), due to the presence ofthe co-propagating probe signal p(t) can be approximated asXPM(f)=H(f)·P(f), where P(f) is the Fourier Transform of the probesignal p(t). In a simple example, the transfer function H(f) can bedefined as a time derivative, which models the effect of changingoptical power level on refractive index, and thus frequency shift.Additional terms can be added to the transfer function H(f), in order tomodel the effects of dispersion, namely; time domain signal distortionsdue to the differing propagation speeds of chirped portions of thatsignal, and “signal walk-off” due to differing propagation speeds ofsignals in different channels. It should be noted that the use of atransfer function H(f) an a mathematical approximation of the XPM effectis known in the art, and thus will not be described in further detailherein.

Preferably, the data signal D(t) transmitted though channel N compriseslive subscriber data traffic. This is advantageous in that use of livetraffic in the monitoring process reduces costs, and minimizes servicedisruptions during the monitoring process. If desired, however, a “testdata” signal designed to emulate live data traffic can be used. Forexample, the “test data” signal may be designed as a pseudo-random bitsequence having a bit rate corresponding to the line rate of channel N.

The probe signal p(t) is preferably designed to generate XPM distortioncomponents in the test signal T(t) that are similar in magnitude tothose that would be generated by live data traffic in channel A. Inprinciple, live data traffic can be used as the probe signal p(t).However, in this case, the probe signal p(t) would need to be measuredat the receiving node 4 b in order to isolate XPM from other distortioneffects. Thus the probe signal p(t) is preferably a predetermined binarysignal (e.g., a simple clock signal) having a known frequency and sharprising and falling edges 22 and 24, as shown in FIG. 2 b. Any arbitraryprobe signal frequency can be used. However, in order to minimize probesignal distortions due to inter-symbol interference and/or dispersion, acomparatively low frequency (e.g., less than about 2.5 GHz) ispreferable. The rising and falling edges 22 and 24 of the probe signalp(t) are selected to be as fast as those of live subscriber datatraffic. With this arrangement, the probe signal p(t) will containhigh-frequency components comparable to those of live subscriber datatraffic, and thus XPM measured using the probe signal p(t) will providea valid approximation of XPM when channel A contains live subscriberdata traffic.

In general, the test signal T(t) arriving at the receiving node 4 bcomprises a distorted version of the original data signal D(t),including distortion components due to noise, dispersion, polarizationdependent effects, and non-linear effects including XPM. The XPM-induceddistortion components comprise chirped portions of the test signal T(t),which have been time-shifted due to dispersion. These XPM-induceddistortion components can be correlated with the received probe signalp(t). Thus the test signal T(t) arriving at the receiving node 46contains a known property (in the form of time-shifted chirped portions)that is uniquely associated with XPM, and can be detected by correlationwith the received probe signal p(t).

Referring back to FIG. 2 a, an XPM monitoring system 26 is located inthe receiving node 4 b, and comprises respective test and probe signalpaths. The test signal path comprises an optical-to-electrical (O/E)converter 28 cascaded with a clock recovery circuit 30. As is known inthe art, these elements respectively operate to convert the receivedtest signal T(t) into a corresponding electrical test signal IT(t), andthen generate a local clock signal that is frequency and phase-lockedwith the electrical test signal IT(t). As is also known in the art,conventional clock recovery circuits use a phase-error signal to huntfor, and maintain, phase lock between a received signal and thegenerated local clock signal. Because of the functional characteristicsof conventional clock recovery circuits, time-shifted portions of thereceived test signal T(t) are carried over into the electrical testsignal IT(t) and produce perturbations 32 within the phase error signal,as shown in FIG. 2 d. Accordingly, the phase error signal generated inthe conventional clock recovery circuit 30 is tapped (at 34) andsupplied to a correlation circuit 36, as shown in FIG. 2 a.

In general, the probe signal path operates to filter the received probesignal p(t), using the previously derived XPM transfer function H(f), inorder to obtain an expected cross phase modulation XPM_(E), as shown inFIG. 2 c. As may be seen from FIGS. 2 c and 2 d, correlation between theexpected cross phase modulation XPM_(E) and the phase error signalenables phase error perturbations 32 corresponding to XPM induceddistortion components of the test signal T(t) to be isolated. Themagnitude of the isolated phase error perturbations 32 can then bedetected in a conventional manner, and is directly indicative ofcross-phase modulation (XPM) between the involved pair of channels.

As may be appreciated, various known methods may be employed to filterthe received probe signal p(t) to obtain the expected cross phasemodulation XPM_(E). In the illustrated embodiment, the probe signal pathcomprises an O/E converter 38, which converts the received probe signalp(t) into a corresponding (analog) electrical probe signal 44. Theelectrical probe signal 44 is then supplied to a high-speedanalog-to-digital converter (ADC) 46, which samples the electrical probesignal 44 at a predetermined sample rate, and generates a correspondingdigital probe signal 48. The sample rate of the ADC 46 is selected toenable complete reconstruction of the original analog electrical probesignal 44. As is known in the art, this can be accomplished by selectingthe sample rate to satisfy the Nyquist criterion for the electricalprobe signal 44. In the present case, because it is important to fullycapture the rising and falling edges 22 and 24 of the electrical probesignal 44, the sample rate is selected to satisfy Nyquist's criterionfor a signal frequency that corresponds to the rise- and fall-time(t_(r)) of the probe signal 16. For example, if t_(r) is selected toemulate 10 GHz data traffic, then the sample rate would be selected tosatisfy Nyquist's criterion for a 10 GHz signal.

The digital probe signal 48 generated by the ADC 46 is then filtered bya digital filter 50 which accepts the XPM transfer function H(f) as aninput. Various known digital filter types may be used to implement thedigital filter 50, such as, for example, Finite Impulse Response (FIR)filters, Infinite Impulse Response (IIR) filters, and Fast FourierTransform (FFT filters). Alternatively, the digital filter 50 may beimplemented using a Random Access Memory Look-up Table (RAM LUT). Ineither case, the output of the digital filter 50 is the (complex)product of the received probe signal p(t) and the XPM transfer functionH(f). A Digital to analog converter (DAC) 56 is coupled to the output ofthe digital filter 50, and operates to convert the filter output into acorresponding analog signal, which is supplied to the correlationcircuit 36.

Because the XPM transfer function H(f) is a linear approximation of XPM(rather than an exact solution), and because the actual magnitude of XPMis unknown, the product of P(f) and H(f) will not exactly model actualXPM distortion components in the test signal T(t). However, it willprovide a provide an expected cross phase modulation XPM_(E) ofsufficient accuracy to permit correlation with the phase error signal.

Accordingly, the phase error signal generated in the conventional clockrecovery circuit is tapped (at 34) and supplied to the correlationcircuit 36, as shown in FIG. 2 a. The correlation circuit 36 alsoreceives the expected cross phase modulation XPM_(E) signal from the DAC56, and operates to extract portions of the phase error signal thatcorrespond with the expected cross phase modulation XPM_(E). Theextracted portions of the phase error signal are then supplied to adetection circuit 40, which measures the power level of the extractedportions of the phase error signal, as a direct indication of XPMbetween channels A and N.

In the embodiment of FIG. 2 a, the correlation circuit 36 is an analogcorrelator, which receives the analog phase error and expected crossphase modulation XPM_(E) signals. However, it will be appreciated that adigital correlator may also be used, in this case, the DAC 56 will beomitted, so that the correlation circuit 36 receives the digital outputof the digital filter 50. An analog to digital converter (not shown)must also be inserted between the clock recovery circuit 30 and thecorrelation circuit 36 in order to convert the analog phase error signalproduced by the clock recovery circuit 30 into a corresponding digitalphase error signal.

A limitation of the embodiment of FIGS. 2 a-2 c is that the receivedprobe signal p(t) contains distortion components, including dispersion,polarization effects, SPM and XPM from any co-propagating signals(including the data signal D(t)) within the link 6. As mentioned above,most of these distortions can be minimized (or in some cases avoided)using known techniques, such as reducing the probe signal frequency.However, in some cases, the probe signal p(t) arriving at the receivingnode 4 b may still be sufficiently distorted to interfere with accuratecorrelation between the expected cross phase modulation XPM_(E) and thephase error signal. It is therefore desirable to filter the receivedprobe signal p(t) to remove at least some of the distortion componentsprior to calculation of the expected cross phase modulation XPM_(E), asis described below with reference to FIG. 3.

As may be appreciated, many techniques may suitably be used to filterremove distortions from the received probe signal p(t). In embodimentsin which the probe signal p(t) is a binary clock signal having a fixed(predetermined) rise and fall time t_(r), a conventional thresholdcomparator-based data regenerator circuit can be used to produce arecovered signal having the required signal shape (principally edgetiming and rise and fall time). The limitation of this approach is thatit cannot easily adjust to changes in the probe signal shape. Forexample, it may be desirable to measure XPM with a range of differentrise and fall times (say, for example, at 6, 10, 14, 18 and 22Picoseconds). A conventional threshold comparator-based data regeneratorcircuit detects rising and falling edges, but is generally insensitiveto rise and fall times per se. Thus a conventional data regeneratorcircuit would fail to provide an accurate version of the original probesignal p(t) to the digital filter 50.

FIG. 3 is a block diagram schematically illustrating principal elementsand operations of a an embodiment of the invention which overcomes thislimitation by filtering the received probe signal p(t) using acompensation filter 42 that compensates at least some of the lineardistortions imposed on the probe signal p(t) by the link 6. The outputof the compensation filter 42 is therefore substantially free of lineardistortion effects, and automatically adjusts to changes in the shape ofthe original probe signal p(t).

As shown in FIG. 3, the digital probe signal 48 generated by the ADC 46is filtered by a digital filter 51 which accepts a compensation functionc(t) as an input. The compensation function c(t), is selected tocompensate at least chromatic dispersion of the link 6. As with thedigital filter 50, various known digital filter types may be used toimplement the digital filter 50, such as, for example, Finite ImpulseResponse (FIR) filters, Infinite Impulse Response (IIR) filters, andFast Fourier Transform (FFT filters). Alternatively, the digital filter51 may be implemented using a Random Access Memory Look-up Table (RAMLUT). In either case, the output of the digital filter 50 is a replicaof the received probe signal p(t), but is substantially free of theeffects of chromatic dispersion.

Various methods may be used to derive the compensation function c(t). Inthe example of FIG. 3, the received probe signal p(t) is monitored (at52) immediately upstream of the O/E converter 38 in order to detectsignal quality parameters indicative of chromatic dispersion in channelA. In preferred embodiments, the signal quality parameters comprise adirect measurement of dispersion at the channel wavelength interest.However, other signal quality parameters (such as, for example, eyeclosure) may be used as a proxy for the dispersion. Any of the signalquality parameters may be detected based on an optical signal obtainedby tapping the received optical probe signal, as shown in FIG. 3. Acompensation function c(t) which optimizes the detected parameters canthen be derived (at 54) deterministically and/or adaptively, using knowntechniques.

It should be noted that the functional step of deriving the compensationfunction c(t) can be implemented by any suitable combination of hardwareand software, which may be co-located with the receiver 46 or any otherlocation. In embodiments in which the detected parameters comprisedirect measurement of dispersion, the compensation function c(t) can becalculated to minimize (and preferably eliminate) the total dispersion.Where eye closure is used as a proxy, then the compensation functionc(t) would be calculated to optimize this value.

The output of the digital filter 50 is a substantially dispersion freereplica of the received probe signal p(t). This signal can then besupplied to the digital filter 50 and filtered using the previouslycalculated XPM transfer function H(f) to obtain the expected cross phasemodulation XPM_(E)[=H(f)·P(f)]. The Digital-to-analog converter (DAC) 56generates a corresponding analog XPM_(E) signal 58, which is thensupplied to the correlation circuit 36 and used to isolate correspondingportions of the phase error signal 34, as described above with referenceto FIGS. 2 a-2 d. As in the embodiment of FIG. 2 a, a digitalcorrelation circuit 36 may also be used in the embodiment of FIG. 3, inwhich case the DAC 56 will be omitted, and an analog to digitalconverter (not shown) inserted between the clock recovery circuit 30 andthe correlation circuit 36.

As described above, the embodiments of FIGS. 2 and 3 provide a directmeasure of XPM in the link 6. However, it does not provide any directinformation concerning self-phase modulation (SPM). One way ofaddressing this limitation relies on the fact that the mechanismsinvolved in producing both XPM and SPM are related, so that SPM can bedescribed as a function of XPM. Consequently, for any particular opticalcommunications system, a look-up table can be defined for estimating SPMfrom the detected XPM. The data used to populate the look-up table may,for example, be based on experimental data obtained during the set-upand commissioning of the optical communications system. If desired, thelook-up table data may be updated, e.g., using fresh experimental dataobtained during maintenance of the optical communications system toaccommodate migration of the optical component performance.

FIGS. 4 a-4 c schematically illustrate principal operations of amonitoring system in accordance with a third embodiment of the presentinvention. Operation of this embodiment is based on the fact that, apartfrom dispersion and polarization effects (both of which can becompensated using known techniques), the received data signal D_(OUT)(t)arriving at the receiving node 4 b through the link 6 contains noise dueto intersymbol interference, ASE and Thermal noise, and phasenon-linearities. Within a sufficiently long “flat region” of thereceived data signal D_(OUT)(t) (i.e., a string of successive binary“1”s or “0”s), variance due to intersymbol interference is avoided, sothat the total noise power will comprise ASE and Thermal noise, andphase non-linearities. As shown in FIG. 4 b, the ASE and Thermal noiseis substantially independent of dispersion, while phase non-linearitiesare strongly dispersion dependent. Accordingly, phase non-linearitiescan be isolated and monitored by detecting changes in the noise power asa function of changing dispersion.

As shown in FIG. 4 a, the present invention provides a variabledispersion unit (VDU) 60, which operates under control of a suitabledispersion control signal Vd to generate the test signal T(t) byimposing a selected amount of dispersion to the received data signalD_(OUT)(t). The test signal T(t) is then supplied to an O/E converter64, which generates a corresponding electrical test signal 66. Theelectrical test signal 66 is then supplied to a pair of detectioncircuits, namely: a flat signal detection circuit 68 which operates todetect a flat region of the electrical test signal 66, in the form of astring of successive bits of equal value; and a noise detector 70, whichoperates to detect a signal parameter indicative of noise. When a flatregion is detected, a noise parameter value at or near the center of theflat region is sampled (or derived) and supplied to a signal processor72. The sampled parameter value can then be stored in a table 74, alongwith the dispersion control signal value Vd, which serves as anindicator of the dispersion imposed by the VDU 60. By progressivelychanging the dispersion control signal value Vd (e.g., periodically, orafter each time a sampled parameter value is stored in the table 74),the signal processor 72 can accumulate data indicative of therelationship between the noise power and dispersion in the table 74.This data can then be analyzed to detect the effects of phasenon-linearities within the link 6, as will described in greater detailbelow with reference to FIG. 5.

FIG. 5 is a block diagram schematically illustrating principal elementsof a monitoring system 76 for implementing the monitoring method ofFIGS. 4 a-c. As shown in FIG. 5, the monitoring system comprises a dataregenerator (known, for example, from U.S. Pat. No. 4,823,360) which hasbeen modified to include the flat signal detection circuit 68, and thesignal processor 72, which includes a Random Access Memory RAM 78 forstoring noise and VDU data. The data regenerator is coupled downstreamof the OLE converter 64, and includes a performance monitor 80 and a setof three parallel data paths, namely: a pair of errored paths whichproduce respective error signals 82 and 84; and an optimum path whichproduces a recovered data signal 86. Each of the data paths includes athreshold comparator 88 cascaded with a retiming circuit 90. Inaddition, the two errored paths include an error counting circuit 92coupled to receive errored data from its respective retiming circuit 90,and recovered data 86 from the optimum path. Together, the errorcounting circuits 92 of the data regenerator provide the functionalityof the noise detection circuit 70 of FIG. 4 a. Outputs of the errorcounting circuits 92 are coupled to the performance monitoring circuit80, which controls operation of the data regenerator. The flat signaldetection circuit 68 is inserted into the (otherwise conventional) dataregenerator, downstream of the retiming circuits 90, as shown in FIG. 5.

Various techniques may be used to implement the flat signal detectioncircuit 68. In the example of FIG. 5, the flat signal detection circuit68 comprises an n-bit shift register 94 connected to the output of eachretiming circuit. Within each of the errored data paths, the respectiven-bit shift register 94 a,b is coupled between the retiming circuit 90and the error counting circuit 92, and acts as a delay line to maintainsynchrony between signals in the three paths. In the optimum data path,an n-port AND-gate 96 is coupled to each bit of the n-bit shift register94 c, such that a binary “1” will appear at the AND-gate output 98 whenat least n successive bits of the recovered data signal 96 have the samevalue. For example, when n successive binary “1”s of the recovered datasignal 86 are latched into the n-bit shift register 94 c, a binary “1”will appear at the AND-gate output 98, indicating that a flat region ofat least n bits in length has been detected. As shown in FIG. 5, theAND-gate output 98 is coupled to an input of the signal processor 72, sothat detection of a flat region can trigger calculation and storage ofnoise and VDU data by the signal processor 72.

As shown in FIG. 5, each of the n-bit shift registers 94 is tapped at abit lying at or near the center of the shift register 94. For example,in FIG. 5, each shift register has a length of 5 bits, and is tapped atthe center (i.e., the third) bit. This means that each error countingcircuit 92 will operate on the basis of the data bit located at (ornear) the center in the sequence of data bits currently stored in theshift register 94. In the case of a flat portion of the recovered datasignal 86 latched into the shift register 94 c, this means that theerror counting circuits 92 will receive the center-bit of the flatportion substantially simultaneously with detection of the flat portionby the n-port AND-gate 96. Noise values calculated by the signalprocessor 72 at a timing of the detection signal from the n-portAND-gate 96, will therefore pertain to the center portion of the flatregion.

The number (n) of bits in each shift register 94 is preferably selectedbased on the number of successive bits (of the received data signal 18)that are subject to inter-symbol interference. With this arrangement,the noise value calculated by the signal processor 72, again, at atiming of the detection signal from the n-port AND-gate 96, will besubstantially free of noise due to inter-symbol interference.

In operation, the electrical test signal 66 generated by the O/Econverter 64 is supplied to a non-inverting (+) input of each of thethreshold comparators 88. The inverting (−) inputs of the thresholdcomparators 88 are connected to receive comparison or threshold levelsV+, Vopt, and V− respectively from the performance monitor 80. Eachthreshold comparator 88 generates a binary signal which reflects theresult of comparison between the electrical test signal and itsrespective threshold level. Threshold levels V+ and V− are respectivelyhigher and lower than the optimum threshold level, Vopt, and producecomparison results containing more errors than Vopt. The binary signalgenerated by each threshold comparator 88 is retimed by a respectiveretiming circuit 90 using a local clock signal (which may, for example,be generated by a conventional clock recovery circuit—not shown in FIG.5). Each retiming circuit 90 (which may be provided by a conventionalD-type flip-flop circuit) generates a respective data signal. The datasignals of the two errored paths are based on the errored binary signalsderived from V+ and V−, and may therefore be referred to as errored datasignals. The retimed binary signal derived from Vopt is an optimumsolution that most nearly reflects the original data content of thereceived test signal, and is used as the recovered data signal 96.

Each of the errored data signals is latched through a respective n-bitshift register 94 a,b of the flat signal detection circuit 68 andsupplied to the respective error counting circuit 92, which alsoreceives the recovered data signal 86. Each error counting circuit 92generates an error signal 82 that is indicative of the differencebetween its errored data signal and the regenerated data signal 86.Typically, the error counting circuits 92 generate error signals whichare proportional to the Bit Error Rate (BER), or, in some cases, thelogarithm of the Bit Error Rate (logBER), of the respective errored datasignal relative to the regenerated data signal. Taken together, the twoerror signals 82 provide an indication of the BER (or logBER) of thereceived test signal as a function of threshold voltage, as shown inFIG. 4 c. This relationship is a direct indicator of noise in the testsignal T(t). Accordingly, as shown in FIG. 5, the two error signals 82and the threshold levels V+ and V− are tapped and supplied to the signalprocessor 72, which uses these values to calculate a noise parametervalue indicative of the noise in the test signal T(t). The noiseparameter value and the current dispersion unit control signal value Vdcan then be saved in the RAM 78. As mentioned previously, byprogressively changing the dispersion control signal value Vd (e.g.,periodically, or after each save operation), the signal processor 72 canaccumulate data indicative of the relationship between the noise andimposed dispersion. This data can then be analyzed to detect the effectsof phase non-linearities within the link 6.

As described in Applicant's co-pending U.S. patent application Ser. No.10/145,035, filed May 15, 2002, some optical networking equipmentutilize high-speed Analog-to-Digital Converters (ADCs) to convertreceived data traffic into digital signals for data recovery and systemmanagement. The sample rate of these ADCs is preferably chosen tosatisfy Nyquist's theorem for the received signal traffic, which meansthat the complete received signal waveform can be recovered from thedigital data stream produced by the ADC. FIG. 6 illustrates a monitoringsystem in accordance with the present invention, which is usable in suchnetworking equipment, and particularly suitable for isolating effects ofXPM and SPM.

As shown in FIG. 6, each channel of the receiving node comprises an O/Econverter 100 which converts the received data signal D_(OUT)(t) into acorresponding electrical data signal. An m-bit ADC 102 (where m ispreferably at least 2, and may be as high as 8 or more) samples theelectrical data signal from the O/E converter 100, at a predeterminedsample rate, and generates a corresponding digital signal 104 in theform of successive digital samples. The digital signal 104 is thensupplied to a decoder circuit 106 for filtering and data recovery using,in the example of FIG. 6, conventional digital equalizer 108 and ForwardError Correction (FEC) 110 circuits. The phase non-linearity monitoringsystem includes a respective channel monitor 112 coupled to eachchannel, and a digital signal processor (DSP) 114 coupled to eachchannel monitor 112 via a data bus 116.

As shown in FIG. 6, each channel monitor 112 comprises a sample memory118 (which may be provided as a conventional random access memory—RAM)for storing sample data in the form of a set of sequential digitalsamples of the digital signal 104. The sample data can then be passed tothe DSP 114 for processing, as will be described in greater detailbelow. Thus the channel monitor 112 utilizes the ADC 102 which operatesnormally to detect the electrical data signal at a timing of the sampleclock. As such, the sample data stored in the sample memory 118 willdirectly reflect the state of the received electrical data signal, andthus the performance of the respective optical channel.

In general, the sample memory 118 may be of any arbitrary size.Preferably, the sample memory 118 will be sized to store samplesgenerated by the ADC 102 within a predetermined interval of time (suchas, for example, in the range of a few tens of microseconds tomilliseconds) and/or encompassing a predetermined number of samples, ora predetermined number of recovered data bits generated by the FECcircuit 110. Samples may be stored continuously, so that the samplememory 118 always contains the most recently generated samples.Preferably, however, the storage operation is controlled by a “write”signal 120 generated by the DSP. For example, the channel monitor 112may be provided with a controller 122 which is responsive to the “write”signal 120 to flush the contents of the sample memory 118, and thenstore a predetermined number of successive samples generated by the ADC102. This arrangement has the advantage that the DSP 114 cansimultaneously control the storage of sample data in every channelmonitor 112 of a multi-channel system. By properly accounting forpropagation delay of the “write” signal 120 between the DSP 114 and eachof the channel monitors 112, it is possible to ensure that the storageoperation is executed substantially simultaneously across all of thechannels of the optical communications system 2. As a result, the sampledata stored by all of the channel monitors 112 will accurately representa “snap shot” of the state of the optical communications system 2 duringthe involved time interval.

As may be appreciated, the simultaneous storage of sample data acrossall of the channels of the optical communications system 2 facilitatescorrelation of sample data from each channel monitor 112, and thushighly accurate analysis of cross-channel effects such as, for example,cross phase modulation (XPM). In particular, the system of FIG. 6 iscapable of implementing the method described above with reference toFIGS. 2 and 3. Thus, a probe signal p(t) can be transmitted (andreceived) though channel-A, while a data signal 18 is transmittedthrough channel-N and received as a test signal 20. In response to the“write” signal from the DSP 114, the respective channel monitors 112will store sample data representing the received probe and test signalsp(t) and T(t). Because the sample rates of the respective ADC's 102satisfy Nyquist's theorem, complete reconstruction of the sampledsignals is possible, allowing the DSP 114 to perform highly accuratecorrelation between the probe signal 16 and distortions in the receivedtest signal 20.

In general, at least some differences in the timing of the storageoperation within each channel monitor 112 must be expected. That is,precise simultaneity across all of the channel monitors 112 will not beachieved. However, provided that there is at least some overlap in thetiming of the storage operation, then sample data from different channelmonitors 112 can be correlated (at least to the extent that the degreeof overlap permits) and cross-channel effects analyzed. The minimumtolerable degree of overlap will generally depend on the minimum amountof signal correlation required to analyze a desired cross-channeleffect.

The transfer of sample data to the digital signal processor (DSP) 114can conveniently be controlled by a “read” signal 124 generated by theDSP 114. This arrangement enables respective sample data from each ofmultiple channel monitors 112 to be transferred to the DSP 114 forfurther analysis. Synchronized storage of sample data within the samplememory 118 of each channel monitor 112 allows indexing of the samples sothat the transfer of sample data to the DSP 114 does not need to bereal-time. Because each channel monitor 112 stores samples representinga substantially simultaneous time intervals across all channels of theoptical communications system 2, information can be transferred fromeach channel monitor 112, in turn, without loss of correlation betweensamples stored by different channel monitors 112.

As may be appreciated, the accuracy with which the DSP 114 can calculateperformance parameters of the communications system 2 is dependent onthe degree to which the sample data stored by each channel monitor 112reflects its respective electrical channel signal. This, in turn, willbe dependent on the resolution of the ADC 102, and the sample frequencyF_(S) of the sample clock. Clearly, increasing the resolution of the ADC102 (e.g., using an 8-bit ADC as opposed to a 2- or 4-bit ADC) increasesprecision of each sample. However, this increased precision is obtainedat a cost of increased expense and complexity.

As shown in FIG. 6, the channel monitor 112 also includes a data memory126 (which may also be provided as a conventional RAM) for storingsequential bits of the recovered digital data stream generated by thedecoder circuit 106. The storage of both sample data and recovereddigital data enables an increased range of signal analysis to beperformed by the DSP 114, including detection of self-phase modulation(SPM), as will be described below.

The sample memory 118 is coupled to the ADC 102 as described above inorder to store successive samples generated by the ADC 102. Similarly,the data memory 126 is coupled to the output of the FEC circuit 110 inorder to store successive data bits of the recovered digital datastream. Because the physical characteristics of the ADC 102 and thedecoder circuit 106 are well characterized, it is possible to controlthe storage operations such that each data bit saved in the data memory126 is properly associated with corresponding digital samples stored inthe sample memory 118. In the illustrated embodiment, this functionalityis implemented by means of a synchronization circuit 128 which operateson the basis of a trigger signal 130 generated by the FEC circuit 110(and indicative of the timing of each corrected bit generated by the FECcircuit 110), in combination with the known propagation delay betweenthe ADC 102 and the output of the FEC circuit 110.

As will be appreciated, the storage of (correlated) sample data andcorrected bits within each channel of the optical communications system2 allows the DSP 114 to analyze the received signal wave form (asrepresented by the sample data) in direct relation to the correspondingdata extracted from that wave-form.

As is known in the art, the effects of SPM vary with both dispersion andthe transmission power level of an optical signal. These effects aregenerally well characterized for the special case of an isolated pulse.Accordingly, for the purposes of detecting the effects of self-phasemodulation (SPM), the stored corrected bits are analyzed to locate a“test signal” in the form of an isolated pulse. For the purposes of thisanalysis, such an isolated pulse may, for example, take the form of oneor more successive binary “1”s preceded and followed by a sufficientnumber of successive binary “0”s to avoid variations due to inter-symbolinterference. Isolated pulses of this type may occur “naturally” withinthe data signal D(t) with sufficient regularity to enable SPMmonitoring. If desired, however, the test signal may also be insertedinto the data signal D(t) at regular intervals. For example, apredetermined bit sequence containing an isolated pulse may be insertedinto the “training sequence” bits that are reserved as part of thestandard transmission format. Once the test signal (isolated pulse) hasbeen detected, the corresponding sample data can be analyzed to evaluatethe associated signal waveform. Changes in the waveform shape withvariations in either dispersion, or the transmission power level of thedata signal D(t) from the sending node 4 a provides a direct indicationof the magnitude of SPM in the link 6.

An alternative approach utilizes the fact that inter-symbol interference(ISI) can be described as being comprised of linear and non-linear ISIcomponents. While both of these components are first-order dispersioneffects, the linear ISI component primarily results from dispersion,filtering etc.; while the non-linear ISI component results,predominantly, from SPM-induced chirp. Accordingly, SPM can be evaluatedby detecting the non-linear ISI component within a “test signal” in theform of the received data signal D_(OUT)(t). As is known in the art,digital equalizers 108 of sufficiently high performance (in terms ofcomprehensive equalization and line rate) will automatically detectnon-linear ISI. Accordingly, in embodiments in which the decoder circuit106 includes a sufficiently comprehensive digital equalizer 108, it ispossible to obtain a direct measure of non-linear ISI by tapping thedigital equalizer 108.

In the embodiment of FIG. 6, detection of the non-linear ISI can also beaccomplished by the DSP 114. In this case, the channel monitor operatesto store temporally-correlated sample data and recovered data bits,which can then be transmitted to the DSP 114, as described above. Thus,the sample data contains sufficient information to enable competereconstruction of the received data signal D_(OUT)(t) waveformcorresponding to the stored data bits. This enables the DSP 114 toprocess the sample data, using known linear equalization techniques, toremove effects of linear ISI. Similarly, known digital signal processingtechniques can be used to process the sample data to remove effects ofdispersion and polarization. Because, the DSP 114 operates “off-line”(that is, out of the path of live data traffic), an inability of the DSP114 to complete these processing steps in real-time is of noconsequence.

Following the above digital signal processing steps (i.e. removal oflinear ISI, dispersion and polarization effects), any remaining ISIwithin the processed sample data will be the non-linear ISI component.This can be evaluated by comparison between the (processed) sample dataand the corresponding recovered data bits, and provides a directindication of SPM.

The embodiment(s) of the invention described above is(are) intended tobe exemplary only. The scope of the invention is therefore intended tobe limited solely by the scope of the appended claims.

It will be appreciated by persons skilled in the art that the presentinvention is not limited to what has been particularly shown anddescribed herein above. In addition, unless mention was made above tothe contrary, it should be noted that all of the accompanying drawingsare not to scale. A variety of modifications and variations are possiblein light of the above teachings without departing from the scope andspirit of the invention, which is limited only by the following claims.

What is claimed is:
 1. A method of monitoring phase nonlinearities of anoptical communications system, the method comprising steps of:generating a test signal using a data signal transmitted through atleast a portion of the optical communications system and received at aselected monitoring point of the optical communications system, the testsignal comprising the received data signal and including a predeterminedproperty that is uniquely associated with at least one phasenonlinearity of the optical communications system; and detecting thepredetermined property of the test signal at the selected monitoringpoint; generating the test signal comprising: receiving the data signalthrough a wavelength channel of the optical communications system, thedata signal including a plurality of flat regions, each flat regionhaving at least a predetermined minimum length; and varying dispersionof the wavelength channel; and the predetermined property comprising avariation in a noise metric of the received data signal that correlateswith a change in the dispersion of the wavelength channel.
 2. A methodas claimed in claim 1, wherein the data signal comprises live datatraffic of the optical communications system.
 3. A method as claimed inclaim 1, wherein each flat region comprises a plurality of successivebits of equal value.
 4. A method as claimed in claim 3, wherein thepredetermined minimum length of each flat region is selected tosubstantially avoid inter-symbol interference within the respective flatregion.
 5. A method as claimed in claim 4, wherein the predeterminedminimum length of each flat region is at least three bits.
 6. A methodas claimed in claim 1, wherein the step of varying dispersion of thewavelength channel comprises a step of successively imposing each one ofa set of predetermined dispersion values on the wavelength channel.
 7. Amethod as claimed in claim 6, wherein the step of detecting thepredetermined property of the test signal comprises, for each imposeddispersion value, steps of: detecting a flat region within the receiveddata signal; measuring the noise metric of the received data signal, ata timing of a center portion of the detected flat region; and storingthe measured noise metric value in association with the imposeddispersion value.
 8. A method as claimed in claim 7, wherein the step ofmeasuring the noise metric comprises a step of measuring any one or moreof: a bit error rate; a signal-to-noise ratio; and an eye-opening ratio.9. An apparatus for monitoring phase nonlinearities of an opticalcommunications system, the apparatus comprising: a receiver configuredto receive a data signal through a wavelength channel of the opticalcommunications system, the data signal comprising a plurality of flatregions, each flat region having at least a predetermined minimumlength; a channel monitor configured to generate a test signalcomprising the received data signal and including a predeterminedproperty that is uniquely associated with at least one phasenonlinearity of the optical communications system, the channel monitorcomprising: a variable dispersion unit configured to vary dispersion ofthe wavelength channel; a flat region detector configured to detect aflat region of the received data signal; a noise detector configured todetect a noise metric of the received data signal; and a SignalProcessor configured to detect the predetermined property of the testsignal, the predetermined property comprising a variation in the noisemetric of the received data signal that correlates with a change in thedispersion of the wavelength channel.